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Data Sheet No. PD 97408A August 18, 2009
IRS26072DSPbF
HIGH AND LOW SIDE DRIVER Features
* * * * * * * * * * * * * Floating channel designed for bootstrap operation Integrated bootstrap diode suitable for Complimentary PWM switching schemes only IRS26072DSPBF is suitable for sinusoidal motor control applications IRS26072DSPBF is NOT recommended for Trapezoidal motor control applications Fully operational to 600 V Tolerant to negative transient voltage, dV/dt immune Gate drive supply range from 10 V to 20 V Under-Voltage lockout for both channels 3.3 V, 5 V, and 15 V input logic compatible Matched propagation delay for both channels Lower di/dt gate driver for better noise immunity Outputs in phase with inputs RoHS compliant
Product Summary
Topology VOFFSET VOUT Io+ & I o- (typical) tON & tOFF (typical) high and low side driver 600 V 10 V - 20 V 200 mA & 350 mA 200 ns
Package Options
Typical Applications
* * * * Motor Control Air Conditioners/ Washing Machines General Purpose Inverters Micro/Mini Inverter Drivers
8-Lead SOIC
Typical Connection Diagram
Up to 600 V
Vcc HIN LIN
Vcc HIN LIN COM
VB HO VS LO IRS26072D TO LOAD
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Table of Contents
Description Simplified Block Diagram Typical Application Diagram Qualification Information Absolute Maximum Ratings Recommended Operating Conditions Static Electrical Characteristics Dynamic Electrical Characteristics Functional Block Diagram Input/Output Pin Equivalent Circuit Diagram Lead Definitions Lead Assignments Application Information and Additional Details Parameter Temperature Trends Package Details Tape and Reel Details Part Marking Information Ordering Information
Page
3 3 4 5 6 6 7 7 8 9 10 10 11 21 25 26 27 28
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IRS26072DSPbF
Description
The IRS26072D is a high voltage, high speed power MOSFET and IGBT driver with independent high and low side referenced output channels. Proprietary HVIC and latch immune CMOS technologies enable ruggedized monolithic construction. Logic inputs are compatible with CMOS or LSTTL outputs, down to 3.3 V. The output drivers feature a high-pulse current buffer stage designed for minimum driver cross-conduction. The floating channel can be used to drive N-channel power MOSFETs or IGBTs in the high side configuration up to 600 V.
Simplified Block Diagram
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Typical Application Diagram
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Qualification Information
Industrial Qualification Level
Comments: This IC has passed JEDEC industrial qualification. IR consumer qualification level is granted by extension of the higher Industrial level. MSL2 , 260C (per IPC/JEDEC J-STD-020) Human Body Model Class 2 (per JEDEC standard JESD22-A114) Class B (per EIA/JEDEC standard EIA/JESD22-A115) Class I, Level A (per JESD78) Yes
Moisture Sensitivity Level
ESD Machine Model IC Latch-Up Test RoHS Compliant
Qualification standards can be found at International Rectifier's web site http://www.irf.com/ Higher qualification ratings may be available should the user have such requirements. Please contact your International Rectifier sales representative for further information.
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Absolute Maximum Ratings
Absolute maximum ratings indicate sustained limits beyond which damage to the device may occur. All voltage parameters are absolute voltages referenced to COM unless otherwise specified. The thermal resistance and power dissipation ratings are measured under board mounted and still air conditions. Symbol VB VS VHO VCC VLO VIN PW HIN dVS/dt PD RthJA TJ TS TL Definition High side floating supply voltage High side floating supply offset voltage High side floating output voltage Low side and logic fixed supply voltage Low side output voltage Logic and analog input voltages High-side input pulse width Allowable offset supply voltage slew rate Thermal resistance, junction to ambient Junction temperature Storage temperature Lead temperature (soldering, 10 seconds) Min. -0.3 VB - 20 VS - 0.3 -0.3 -0.3 -0.3 500 -- -- -- -- -50 -- Max. 620 VB + 0.3 VB + 0.3 20 VCC + 0.3 VCC + 0.3 -- 50 0.625 200 150 150 300 Units
V
ns V/ns W C/W C
All supplies are fully tested at 25 V. An internal 20 V clamp exists for each supply.
Recommended Operating Conditions
For proper operation, the device should be used within the recommended conditions. All voltage parameters are absolute voltages referenced to COM unless otherwise specified. The VS offset ratings are tested with all supplies biased at 15 V. Symbol VB VS VS(t) VHO VCC VLO VIN TA Definition High side floating supply voltage Static high side floating supply offset voltage Transient high side floating supply offset voltage High side floating output voltage Low side and logic fixed supply voltage Low side output voltage Logic input voltage Ambient temperature Min. VS +10 -8 -50 VS 10 0 0 -40 Max. VS + 20 600 600 VB 20 VCC VCC 125 Units
Logic operation for VS of -8 V to 600 V. Logic state held for VS of -8 V to -VBS. Operational for transient negative VS of -50 V with a 50 ns pulse width. Guaranteed by design. Refer to the Application Information section of this datasheet for more details.
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Package power dissipation @ TA
+25 C
V
C
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Static Electrical Characteristics
(VCC-COM) = (VB-VS) = 15 V and TA = 25 C unless otherwise specified. The VIN and IIN parameters are referenced to COM. The VO and IO parameters are referenced to COM and VS and are applicable to the output leads LO and HO respectively. The VCCUV and VBSUV parameters are referenced to COM and VS respectively. Symbol Definition Min. Typ. Max. Units Test Conditions
o
VIH Logic "1" input voltage 2.5 -- -- VIL Logic "0" input voltage -- -- 0.8 VIN,TH+ Input positive going threshold -- 1.9 -- VIN,THInput negative going threshold -- 1 -- VOH High level output voltage -- 0.8 1.4 IO = 20 mA V VOL Low level output voltage -- 0.2 0.6 VCCUV+ VCC and VBS supply under-voltage positive 8.0 8.9 9.8 VBSUV+ going threshold VCCUVVCC and VBS supply under-voltage negative 6.9 7.7 8.5 VBSUVgoing threshold VCCUVH VCC and VBS supply under-voltage hysteresis 0.35 1.2 -- VBSUVH ILK Offset supply leakage current -- 1 50 VB =VS = 600 V A IQBS Quiescent VBS supply current -- 45 70 VIN = 0 V or 5 V IQCC Quiescent VCC supply current -- 1.1 1.8 mA IIN+ Logic "1" input bias current -- 5 20 VIN = 5 V A IINLogic "0" input bias current -- -- 2 VIN = 0 V Io+ Output high short circuit pulsed current 120 200 -- VO = 0 V or 15 V mA PW 10 s IoOutput low short circuit pulsed current 250 350 -- RBS -- 200 -- Bootstrap resistance Integrated bootstrap diode is suitable for Complimentary PWM schemes only. IRS26072D is suitable for sinusoidal motor control applications. IRS26072D is NOT recommended for Trapezoidal motor control applications. Refer to the Integrated Bootstrap Functionality section of this datasheet for more details.
Dynamic Electrical Characteristics
VCC = VB = 15 V, VS = COM, TA = 25 C and CL = 1000 pF unless otherwise specified. Symbol ton toff tr tf MT PM Definition Turn-on propagation delay Turn-off propagation delay Turn-on rise time Turn-off fall time ton, toff propagation delay matching time PW pulse width distortion Min. 100 100 -- -- -- -- Typ. 200 200 150 50 -- -- Max. 300 300 220 80 50 75 Units Test Conditions
o
ns
VIN = 0V and 5V
PW input =10s
PM is defined as PW IN - PW OUT.
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Functional Block Diagram
VB UV DETECT R HV LEVEL SHIFTER HIN PULSE GENERATOR PULSE FILTER R S Q HO
VS
Integrated BS DIODE
VCC
UV DETECT LO
LIN
DELAY
COM
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Input/Output Pin Equivalent Circuit Diagrams
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Lead Definitions
Symbol VCC VB VS HIN LIN HO LO COM Description Low side and logic power supply High side floating power supply High side floating supply return Logic input for high side gate driver output HO, input is in-phase with output Logic input for low side gate driver output LO, input is in-phase with output High side gate driver output Low side gate driver output Low side supply return
Lead Assignments
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Application Information and Additional Details
* * * * * * * * * * * IGBT/MOSFET Gate Drive Switching and Timing Relationships Matched Propagation Delays Input Logic Compatibility Under-Voltage Lockout Protection Truth Table: Under-Voltage lockout Integrated Bootstrap Functionality Bootstrap Power Supply Design Tolerant to Negative VS Transients PCB Layout Tips Additional Documentation
IGBT/MOSFET Gate Drive The IRS26072D HVIC is designed to drive high side and low side MOSFET or IGBT power devices. Figures 1 and 2 show the definition of some of the relevant parameters associated with the gate driver output functionality. The output current that drives the gate of the external power switches is defined as IO. The output voltage that drives the gate of the external power switches is defined as VHO for the high side and VLO for the low side; this parameter is sometimes generically called VOUT and in this case the high side and low side output voltages are not differentiated.
VB (or VCC)
VB (or VCC)
IO+
HO (or LO) + HO (or LO)
VHO (or VLO)
VS (or COM) VS (or COM )
IO -
Figure 1: HVIC sourcing current
Figure 2: HVIC sinking current
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Switching and Timing Relationships The relationship between the input and output signals of the IRS26072D HVIC is shown in Figure 3. The definitions of some of the relevant parameters associated with the gate driver input to output transmission are given.
LIN or HIN
50% PWIN
50%
t ON
tR 90% 10%
PWOUT
tOFF 90%
tF
LO or HO
10%
Figure 3: Switching time waveforms
During interval A of Figure 4 the HVIC receives the command to turn on both the high and low side switches at the same time; correspondingly, the high and low side signals HO and LO turn on simultaneously.
Figure 4: Input/output timing diagram
Matched Propagation Delays The IRS26072D HVIC is designed for propagation delay matching. With this feature, the input to output propagation delays tON, tOFF are the same for the low side and the high side channels; the maximum difference being specified by the delay matching parameter MT as defined in Figure 6.
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Figure 6: Delay Matching Waveform Definition
Input Logic Compatibility The IRS26072D HVIC is designed with inputs compatible with standard CMOS and TTL outputs with 3.3 V and 5 V logic level signals. Figure 7 shows how an input signal is logically interpreted.
Figure 7: HIN & LIN input thresholds
Under-Voltage Lockout Protection The IRS26072D HVIC provides under-voltage lockout protection on both the VCC low side and logic fixed power supply and the VBS high side floating power supply. Figure 8 illustrates this concept by considering the VCC (or VBS) plotted over time: as the waveform crosses the UVLO threshold, the under-voltage protection is entered or exited. Upon power up, should the VCC voltage fail to reach the VCCUV+ threshold, the gate driver outputs LO and HO will remain disabled. Additionally, if the VCC voltage decreases below the VCCUV- threshold during normal operation, the under-voltage lockout circuitry will shutdown the gate driver outputs LO and HO. Upon power up, should the VBS voltage fail to reach the VBSUV threshold, the gate driver output HO will remain disabled. Additionally, if the VBS voltage decreases below the VBSUV threshold during normal operation, the undervoltage lockout circuitry will shutdown the high side gate driver output HO.
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The UVLO protection ensures that the HVIC drives external power devices only with a gate supply voltage sufficient to fully enhance them. Without this protection, the gates of the external power switches could be driven with a low voltage, which would result in power switches conducting current while with a high channel impedance, which would produce very high conduction losses possibly leading to power device failure.
VCC (or V BS ) V CCUV + ( or V BSUV + )
VCCUV (or V BSUV - )
Time UVLO Protection ( Gate Driver Outputs Disabled ) Normal Operation Normal Operation
Figure 8: UVLO protection
Truth Table: Under-Voltage lockout Table 2 provides the truth table for the IRS26072D HVIC. The 1 line shows that for VCC below the UVLO threshold both the gate driver outputs LO and HO are disabled. After VCC returns above VCCUV, the gate driver outputs return functional. The 2 line shows that for VBS below the UVLO threshold, the gate driver output HO is disabled. After VBS returns above VBSUV, HO remains low until a new rising transition of HIN is received. The last line shows the normal operation of the HVIC.
nd st
VCC UVLO VCC UVLO VBS Normal operation VBS
outputs LO 0 LIN LIN HO 0 0 HIN
Table 2: UVLO truth table
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Integrated Bootstrap Functionality The IRS26072D HVIC embeds an integrated bootstrap FET that eliminates the need of external bootstrap diodes and resistors allowing an alternative drive of the bootstrap supply for a wide range of applications. A bootstrap FET is connected between the high side floating power supply VB and the low side and logic fixed power supply VCC (see Fig. 9).
VCC
Bootstrap FET
VB
Figure 9: Simplified Bootstrap FET connection The bootstrap FET is suitable for complimentary PWM switching schemes only. Complimentary PWM refers to PWM schemes where the HIN & LIN input signals are alternately switched on and off. IRS26072D is suitable for sinusoidal motor control and the integrated bootstrap feature can be used either in parallel with the external bootstrap network (diode and resistor) or as a replacement of it. The use of the integrated bootstrap as a replacement of the external bootstrap network may have some limitations at very high PWM duty cycle, corresponding to very short LIN pulses, due to the bootstrap FET equivalent resistance RBS. IRS26072D is NOT recommended for trapezoidal motor control, even if an external bootstrap network is employed in parallel. The bootstrap FET is conditioned as follows: * bootstrap turns-off (immediately) or stays off when either: HO goes/stays high; VB goes/ stays high (> 1.1*VCC); * bootstrap turns-on when: LO is high (low side is on) AND VB is low (<1.1*VCC); LO and HO are low after a transition of LIN from high to low AND VB goes low (<1.1*VCC) before a fixed time of 20us; LO and HO are low after a transition of HIN from high to low AND VB goes low (<1.1*VCC) before a re-triggerable time of 20us. In this case the time counter is kept in reset state until VB goes high (>1.1VCC). In Figure 10 the BootFET timing diagram details are represented.
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20 us timer Timer is reset counter Timer is reset Timer expired
HIN
LIN
BootStrap Fet
VB 1.1*Vcc
+ -
Figure 10: BootFET timing diagram
Bootstrap Power Supply Design For information related to the design of the bootstrap power supply while using the integrated bootstrap functionality of the IRS26072D, please refer to Application Note 1123 (AN-1123) entitled "Bootstrap Network Analysis: Focusing on the Integrated Bootstrap Functionality." This application note is available at www.irf.com. For information related to the design of a standard bootstrap power supply (i.e., using an external discrete diode) please refer to Design Tip 04-4 (DT04-4) entitled "Using Monolithic High Voltage Gate Drivers." This design tip is available at www.irf.com.
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Tolerant to Negative VS Transients A common problem in today's high-power switching converters is the transient response of the switch node's voltage as the power devices switch on and off quickly while carrying a large current. A typical 3-phase inverter circuit is shown in Figure 11; where we define the power switches and diodes of the inverter. If the high-side switch (e.g., the IGBT Q1 in Figures 12 and 13) switches off, while the U phase current is flowing to an inductive load, a current commutation occurs from high-side switch (Q1) to the diode (D2) in parallel with the low-side switch of the same inverter leg. At the same instance, the voltage node VS1, swings from the positive DC bus voltage to the negative DC bus voltage.
Figure 11: Three phase inverter
DC+ BUS
Q1 ON IU VS1 D2
Q2 OFF
DC- BUS
Figure 12: Q1 conducting
Figure 13: D2 conducting
Also when the V phase current flows from the inductive load back to the inverter (see Figures 14 and 15), and Q4 IGBT switches on, the current commutation occurs from D3 to Q4. At the same instance, the voltage node, VS2, swings from the positive DC bus voltage to the negative DC bus voltage.
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Figure 14: D3 conducting
Figure 15: Q4 conducting
However, in a real inverter circuit, the VS voltage swing does not stop at the level of the negative DC bus, rather it swings below the level of the negative DC bus. This undershoot voltage is called "negative VS transient". The circuit shown in Figure 16 depicts one leg of the three phase inverter; Figures 17 and 18 show a simplified illustration of the commutation of the current between Q1 and D2. The parasitic inductances in the power circuit from the die bonding to the PCB tracks are lumped together in LC and LE for each IGBT. When the high-side switch is on, VS1 is below the DC+ voltage by the voltage drops associated with the power switch and the parasitic elements of the circuit. When the high-side power switch turns off, the load current momentarily flows in the lowside freewheeling diode due to the inductive load connected to VS1 (the load is not shown in these figures). This current flows from the DC- bus (which is connected to the COM pin of the HVIC) to the load and a negative voltage between VS1 and the DC- Bus is induced (i.e., the COM pin of the HVIC is at a higher potential than the VS pin).
Figure 16: Parasitic Elements
Figure 17: VS positive
Figure 18: VS negative
In a typical motor drive system, dV/dt is typically designed to be in the range of 3-5 V/ns. The negative VS transient voltage can exceed this range during some events such as short circuit and over-current shutdown, when di/dt is greater than in normal operation. International Rectifier's HVICs have been designed for the robustness required in many of today's demanding applications. An indication of the IRS26072D's robustness can be seen in Figure 19, where there is represented the IRS26072D Safe Operating Area at VBS=15V based on repetitive negative VS spikes. A negative VS transient voltage falling in the grey area (outside SOA) may lead to IC permanent damage; vice versa unwanted functional anomalies or permanent damage to the IC do not appear if negative Vs transients fall inside SOA.
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At VBS=15V in case of -VS transients greater than -16.5 V for a period of time greater than 50 ns; the HVIC will hold by design the high-side outputs in the off state for 4.5 s.
Figure 19: Negative VS transient SOA @ VBS=15V Even though the IRS26072D has been shown able to handle these large negative VS transient conditions, it is highly recommended that the circuit designer always limit the negative VS transients as much as possible by careful PCB layout and component use.
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PCB Layout Tips Distance between high and low voltage components: It's strongly recommended to place the components tied to the floating voltage pins (VB and VS) near the respective high voltage portions of the device. Please see the Case Outline information in this datasheet for the details. Ground Plane: In order to minimize noise coupling, the ground plane should not be placed under or near the high voltage floating side. Gate Drive Loops: Current loops behave like antennas and are able to receive and transmit EM noise (see Figure 20). In order to reduce the EM coupling and improve the power switch turn on/off performance, the gate drive loops must be reduced as much as possible. Moreover, current can be injected inside the gate drive loop via the IGBT collector-to-gate parasitic capacitance. The parasitic auto-inductance of the gate loop contributes to developing a voltage across the gate-emitter, thus increasing the possibility of a self turn-on effect.
VB (or VCC ) HO (or LO ) VS (or COM)
Figure 20: Antenna Loops
RG
IGC
CGC
Gate Drive Loop
VGE
Supply Capacitor: It is recommended to place a bypass capacitor between the VCC and COM pins. This connection is shown in Figure 21. A ceramic 1 F ceramic capacitor is suitable for most applications. This component should be placed as close as possible to the pins in order to reduce parasitic elements.
Up to 600V Vcc HIN1,2,3 LIN1,2,3 Vcc HIN1,2,3 LIN1,2,3 VB1,2,3 HO1,2,3 VS 1,2,3 TO LOAD LO1,2,3 COM
GND
Figure 21: Supply capacitor
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Routing and Placement: Power stage PCB parasitic elements can contribute to large negative voltage transients at the switch node; it is recommended to limit the phase voltage negative transients. In order to avoid such conditions, it is recommended to 1) minimize the high-side source to low-side collector distance, and 2) minimize the low-side emitter to negative bus rail stray inductance. However, where negative VS spikes remain excessive, further steps may be taken to reduce the spike. This includes placing a resistor (5 or less) between the VS pin and the switch node (see Figure 22), and in some cases using a clamping diode between COM and VS (see Figure 23). See DT04-4 at www.irf.com for more detailed information.
DC+ BUS DC+ BUS
VB C BS HO
VB C BS HO
VS
R VS
LO
To Load
VS
RVS D VS
LO
To Load
COM
COM
DC- BUS
DC- BUS
Figure 22: VS resistor
Figure 23: VS clamping diode
Additional Documentation Several technical documents related to the use of HVICs are available at www.irf.com; use the Site Search function and the document number to quickly locate them. Below is a short list of some of these documents. DT97-3: Managing Transients in Control IC Driven Power Stages AN-1123: Bootstrap Network Analysis: Focusing on the Integrated Bootstrap Functionality DT04-4: Using Monolithic High Voltage Gate Drivers AN-978: HV Floating MOS-Gate Driver ICs
Parameter Temperature Trends
Figures 24-41 provide information on the experimental performance of the IRS26072D HVIC. The line plotted in each figure is generated from actual experimental data. A small number of individual samples were tested at three temperatures (-40 C, 25 C, and 125 C) in order to generate the experimental curve. The line labeled Exp. consist of three data points (one data point at each of the tested temperatures) that have been connected together to illustrate the understood temperature trend. The individual data points on the curve were determined by calculating the averaged experimental value of the parameter (for a given temperature).
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800 700 600 500
Exp.
800 700 600
Exp.
tOFF (ns)
500 400 300 200 100 0 -50
tON (ns)
400 300 200 100 0 -50 -25 0 25 50 75 100 125
-25
0
25
50
75
100
125
Temperature (o C)
Temperature (o C)
Fig. 24. Turn-on Propagation Delay vs. Temperature
200 180 160 140
Fig. 25. Turn-off Propagation Delay vs. Temperature
60 50 40
tR (ns)
120 100 80
Exp.
tF (ns)
30 20 10 0 -50
Exp.
60 40 20 0 -50
-25
0
25
50
75
100
125
-25
0
25
50
75
100
125
Temperature (oC)
Temperature (oC)
Fig. 26. Turn-on Rise Time vs. Temperature
2.5 450 400 2.0
Exp.
Fig.27. Turn-off Fall Time vs. Temperature
350
VOL_LO1 (mV)
LIN1_VTH- (V)
300 250 200 150 100 50
Exp.
1.5
1.0
0.5
0.0 -50
0 -25 0 25 50 75 100 125
-50
-25
0
25
50
75
100
125
Temperature (oC)
Temperature (oC)
Fig. 28. Input Negative Going Threshold vs. Temperature
Fig. 29. Low Level Output Voltage vs. Temperature
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60
12
50
10
ileak1_VCCMAX (A)
40
8
Exp.
30
I QCC1 (mA)
Exp.
6
20 10
4
2
0
-50
-25
0
25
50
75
100
125
0 -50
-25
0
25
50
75
100
125
Temperature (oC)
Temperature (oC)
Fig. 30. Offset Supply Leakage Current vs. Temperature
7 6
Exp.
Fig. 31. Quiescent VCC Supply Current vs. Temperature
80 70 60
5
I QCC0 (mA)
4 3 2 1 0 -50
IQBS10 (A)
50 40 30 20 10 0 -50
Exp.
-25
0
25
50
75
100
125
-25
0
25
50
75
100
125
Temperature (oC)
Temperature (oC)
Fig. 32. Quiescent VCC Supply Current vs. Temperature
80 70 60
Fig. 33. Quiescent VBS Supply Current vs. Temperature
9.6 9.4 9.2 9.0
IQBS11 (A)
VCCUV- (V)
50 40 30 20 10 0 -50
Exp.
8.8 8.6
Exp.
8.4 8.2 8.0 7.8 -50
-25
0
25
50
75
100
125
-25
0
25
50
75
100
125
Temperature (oC)
Temperature (oC)
Fig. 34. Quiescent VBS Supply Current vs. Temperature
Fig. 35. VCC Supply Under-voltage Negative Going Threshold vs. Temperature (c) 2009 International Rectifier
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9.8 9.6 9.4
9.0
8.5
VCCUV+ (V)
VBSUV- (V)
9.2 9.0 8.8 8.6
Exp.
8.0
Exp.
7.5
7.0
6.5
8.4 8.2 -50
6.0 -50
-25
0
25
50
75
100
125
-25
0
25
50
75
100
125
Temperature (oC)
Temperature (oC)
Fig. 36. VCC Supply Under-voltage Positive Going Threshold vs. Temperature
9.5 9.0 8.5
Exp.
Fig. 37. VBS Supply Under-voltage Negative Going Threshold vs. Temperature
0 -50 -50 -100 -25 0 25 50 75 100 125
V BSUV+ (V)
-150
IO+ (mA)
8.0 7.5 7.0 6.5 6.0 -50
-200 -250 -300 -350 -400
Exp. p.
-25
0
25
50
75
100
125
-450
Temperature (oC)
Temperature (oC)
Fig. 38. VBS Supply Under-voltage Positive Going Threshold vs. Temperature
706 606 506
Exp.
Fig. 39. Output High Short Circuit Pulsed Current vs. Temperature
0 -50 -2 -25 0 25 50 75 100 125
Vs1_RST_domin (V)
-4 -6 -8 -10 -12
Exp.
I O- (mA)
406 306 206 106 6 -50
-25
0
25
50
75
100
125
-14
Temperature (o C)
Temperature (o C)
Fig. 40. Output Low Short Circuit Pulsed Current vs. Temperature www.irf.com
Fig. 41. Max -Vs vs. Temperature
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Package Details
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Tape and Reel Details
LOADED TAPE FEED DIRECTION
B
A
H
D F C
NOTE : CONTROLLING DIM ENSION IN M M
E G
CARRIER TAPE DIMENSION FOR Metric Code Min Max A 7.90 8.10 B 3.90 4.10 C 11.70 12.30 D 5.45 5.55 E 6.30 6.50 F 5.10 5.30 G 1.50 n/a H 1.50 1.60
8SOICN Imperial Min Max 0.311 0.318 0.153 0.161 0.46 0.484 0.214 0.218 0.248 0.255 0.200 0.208 0.059 n/a 0.059 0.062
F
D C E B A
G
H
REEL DIMENSIONS FOR 8SOICN Metric Code Min Max A 329.60 330.25 B 20.95 21.45 C 12.80 13.20 D 1.95 2.45 E 98.00 102.00 F n/a 18.40 G 14.50 17.10 H 12.40 14.40
Imperial Min Max 12.976 13.001 0.824 0.844 0.503 0.519 0.767 0.096 3.858 4.015 n/a 0.724 0.570 0.673 0.488 0.566
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Part Marking Information
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Ordering Information
Standard Pack Base Part Number Package Type Form IRS26072D SOIC 8 Tube/Bulk Tape and Reel Quantity XXX XXX IRS26072D SPBF IRS26072D STRPBF Complete Part Number
The information provided in this document is believed to be accurate and reliable. However, International Rectifier assumes no responsibility for the consequences of the use of this information. International Rectifier assumes no responsibility for any infringement of patents or of other rights of third parties which may result from the use of this information. No license is granted by implication or otherwise under any patent or patent rights of International Rectifier. The specifications mentioned in this document are subject to change without notice. This document supersedes and replaces all information previously supplied.
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